1. Field of the Invention
This invention relates to an interference signal removal system for removing a narrow-band interference signal from a received signal that contains a narrow-band interference signal and a desired broadband signal and varies periodically in received power level, and particularly to such a system enabling efficient interference signal removal.
2. Description of the Prior Art
A signal received by a receiver may, for example, contain not only the signal to be received (desired signal) but also an accompanying signal that interferes with the desired signal (interference signal).
An explanation will be given regarding broadband desired signals and narrow-band interference signals, taking the wireless LAN (Local Area Network) of IEEE 802.11 as an example.
It should be noted that the meanings of the terms “broadband” and “narrow-band” are relative. Specifically, a signal whose exclusive bandwidth is considerably broad in comparison with the exclusive bandwidth of a narrow-band interference signal is called a “broadband signal.” For example, a signal whose exclusive bandwidth is ten or more times greater than the exclusive bandwidth of a narrow-band interference signal is called a “broadband signal.” In the case of the wireless LAN under discussion, the exclusive bandwidth of the broadband signal is, for instance, 26 MHz (frequency per wave) and the exclusive bandwidth of the narrow-band signal is, for instance, 2 MHz (frequency per wave).
The IEEE 802.11 wireless LAN is broadly divided into two system types: the DSSS (Direct Sequence Spread Spectrum) system and FHSS (Frequency Hopping Spread Spectrum) system. From the difference in their modulated waves, the signal according to DSSS can be viewed as a broadband signal and the signal according to FHSS can be viewed as a narrow-band signal. Since the two systems conduct wireless communication using the same frequency band and are allowed to interfere systematically, the signals naturally interfere with each other.
DSSS is a communications system that spectrum-spreads and transmits a narrow-band signal as a broadband signal and on the receive side restores the signal to the original narrow-band signal in a demodulation process. In DSSS, therefore, the narrow-band interference signal contained in the received signal can be suppressed by spreading it over the broadband signal in the demodulation process. The ratio between before and after spreading is called the “spreading rate.” When the spreading rate is 128, for example, a gain of about 21 dB (10 LOG 128 to be exact) is obtained.
FHSS is a system that utilizes a broadband for communications by changing the transmit frequency of a narrow-band signal at regular time intervals. When the FHSS system is observed at a fixed time point, therefore, its exclusive bandwidth is narrow, i.e., 2 MHz, and the power of the DSSS of the band is relatively low. As a result, the effect of interference can be suppressed by a receive filter of a receiver utilizing the FHSS system.
In FHSS, interference between signals transmitted by different transmitters using different hopping patterns is unlikely to cause a problem because the probability of the same frequency being used at the same time is low. Moreover, FHSS can frequency hop over a broader band than the DSSS frequency band. Even when strong interference is experienced from DSSS, therefore, signal reception is still possible by using a frequency band where interference is not encountered.
With DSSS, however, the spreading rate may be lowered to increase the signal transfer speed. When the spreading rate is lowered to 11, for example, gain falls to about 10 dB (10 LOG 11 to be exact). When the spreading rate is lowered even more, gain falls still further. It may therefore become impossible to obtain an interference signal suppressing effect.
In addition, Bluetooth (short-range mobile service) and other standards that utilize FHSS are coming into wide use as wireless interfaces. The probability of DSSS signals encountering interference is therefore increasing.
Other instances in which interference may become a problem include interference occurring in adjacent frequency bands between W-CDMA (Wideband-Code Division Multiple Access) communication signals and PHS (Personal Hand phone System) communication signals, interference occurring between 2.4 GHz band wireless LAN (IEEE 802.11) broadband signals and Bluetooth narrow-band signals, interference occurring owing to frequency band sharing between CDMA communication signals and TDMA (Time Division Multiple Access) or FDMA (Frequency Division Multiple Access) communication signals, and interference caused by unpredictable extraneous waves.
Known techniques for removing such inference under consideration include a removal method using an adaptive algorithm and an interference signal removal method using a filter such as a notch filter. One example is the technique for removing narrow-band signals interfering with broadband signals with a notch filter using a multi-rate filter bank described in Application of Complex Multi-rate Filter Bank to a Frequency Band Sharing DS-CDMA/TDMA Signal Blanket Receiver (Electronics and Communications Engineering Journal B-II Vol. J80-BII No 12 December 1997).
An example of a conventional interference signal removal system will now be explained. The interference signal removal system is, for example, installed in a receiver for wireless communications and used to remove interference signals contained in signals received by the receiver.
As exemplified in FIG. 10, the interference signal removal system is equipped with an interference signal estimator 31, an interference signal extractor 32 and a synthesizer 33. The symbol t represents time.
A received signal r(t), which contains a desired broadband signal synthesized with multiple narrow-band interference signals, and a received signal e(t) after removal of interference are input to the interference signal estimator 31. The interference signal estimator 31 estimates interference signals contained in the received signal r(t) using an ordinary adaptive algorithm and outputs an interference signal estimation coefficient h(t+1) based on the estimation result to the interference signal extractor 32.
The received signal r(t) and the interference signal estimation coefficient h(t+1) are input to the interference signal extractor 32. The interference signal extractor 32 extracts a (presumed) interference signal V(t) from the received signal r(t) based on the interference signal estimation coefficient h(t+1) received from the interference signal estimator 31 and outputs the interference signal V(t) to the synthesizer 33.
The synthesizer 33 synthesizes the received signal r(t) and the interference signal V(t) from the interference signal extractor 32 in opposite phase (i.e., so that the interference signal V(t) is removed from the received signal r(t)) and outputs the received signal e(t) removed of the interference signal V(t). Part of the interference-reduced received signal e(t) output by the synthesizer 33 is sent to the interference signal estimator 31 and used for interference signal estimation.
CDMA and an interference signal removal system for CDMA will now be explained.
In a mobile communications system using DS-CDMA, for example, multiplex communication between multiple mobile stations and a base station is achieved by assigning different spreading codes to the individual mobile stations. Specifically, each mobile station uses the spreading code assigned to it to modulate (spread) the signal to be transmitted and the base station demodulates the signals from the individual desired mobile stations by using their assigned spreading codes to despread the received signals. Each mobile station uses its assigned spreading code to despread signals received from the base station and thereby demodulate the signals addressed to itself.
FIG. 11 shows an example of a spreading code sequence composed of, for example, a PN (pseudonoise signal) sequence.
As illustrated, one spreading code unit (for one symbol) is composed of chip data (e.g., a row of “1” values and “−1” values) and a plurality of different spreading codes are generated by varying the pattern of the row of chip data. A characteristic of the spreading codes is that when a given spreading code is shifted by one or more chip times, it loses its correlation with that spreading code.
The drawing indicates one chip data time width (chip interval Tc) and the spreading code time width for one symbol (bit interval T). The time width of the spreading code for one symbol corresponds to the time width of the transmission data (e.g., “1” values and “0” values) transmitted from a transmitter (e.g., a mobile station or a base station) to a receiver (e.g., a base station or a mobile station). In other words, the change rate of the chip data constituting the spreading code is very fast compared with the switching rate of the transmission data spread-modulated by the spreading code (the symbol switching rate).
As explained above, in this type of wireless communication, interference may occur within the broad frequency band whose frequencies are approved and used for communication, owing to unintentional mixing therewith of other (i.e., other than CDMA) narrow-band signals and the like. When the signals of this type exceed the level of interference by noise and the like envisioned at the time of system design, the number of bit errors increases and the quality of receiver reception is markedly degraded.
Another conceivable way to realize the foregoing purpose of efficient frequency band utilization is to achieve multiplex communication using a system like CDMA that communicates using a relatively broad frequency band together with a system like FM (frequency modulation) that communicates using a narrow-band. As a matter of operating principle, it is, for example, possible to achieve effective frequency band utilization by multiplexing an FM or other such analog communication signal with the frequency band of a CDMA spread signal. Unless the CDMA receiver can remove the FM signal etc. from the received signal, however, it and the spread signal will interfere with each other to increase the number of bit errors and degrade the reception quality.
FIG. 12 shows an example of the spectrum of a received signal containing a CDMA spread signal and an FM signal (FM interference wave), in which frequency is represented on the horizontal axis and spectral intensity on the vertical axis.
As an example of a conventional interference signal removal system (interference removal circuit), the one taught by published Japanese Patent Application No. 11-197296 will now be explained with reference to FIGS. 13 to 17. The interference signal removal system taught by this application is intended for use in a base station, mobile station or relay station utilizing CDMA to process a received signal containing a CDMA spread-modulated broadband signal and a narrow-band interference signal so as to remove the interference signal from the received signal or from the I component and Q component of the received signal, and is particularly adapted to utilize the characteristics of the spread signal for interference signal removal.
An example of the interference signal removal system configuration is shown in FIG. 13. The illustrated interference signal removal system processes an input received signal (received signal r(t)) containing a CDMA signal (desired signal) and an FM signal (interference signal) to remove the FM signal from the received signal r(t). In this interference signal removal system, removal of the interference signal from the received signal containing the CDMA signal (CDMA spread-modulated signal) and the interference signal is effected by imparting a time difference of one or more spreading code chip times between two signals into which the received signal is split by a time difference imparting means 41, sending the two signals to extraction means 42, 44 that extract a signal component having correlation between the two signals imparted with the time difference as an interference signal component, and sending the extracted interference signal component to a removal means 43 that uses it to remove the interference signal component from the received signal.
More specifically, the illustrated interference signal removal system comprises a delay device 41 for delaying the received signal, an adaptive filter 42 for extracting an interference signal component from the delayed received signal in accordance with a tap coefficient control signal from a filter tap coefficient arithmetic and control unit 44, a subtractor 43 for removing the interference signal component from the received signal, and the filter tap coefficient arithmetic and control unit 44 for outputting to the adaptive filter 42 the tap coefficient control signal based on the output signal from the subtractor 43 and the delayed received signal.
The configuration and operation of the illustrated circuit will now be explained.
The circuit is supplied with a signal r(t) received by a receiver. The received signal r(t) includes a spread signal, e.g. a CDMA spread-modulated signal, and a narrow-band communication signal constituting an interference signal (e.g., an FM-modulated signal). The symbol t represents time and, in the present case, is an integer discrete value representing the minimum unit of one sample time.
The input signal r(t) is first split into two signals, one of which is input to the delay device 41 and the other of which is input to the subtractor 43.
The delay device 41 functions to delay the input signal by one or more spreading code chip time widths. This time difference can be preset to a value that, for example, is sufficient to eliminate the spread signal correlation component between the two signals but allow the correlation component of the interference signal to be removed to remain.
Specifically, the signal output by the delay device 41 is represented by r(t−τ), where τ is the delay time imparted by the delay device 41.
The signal r(t−τ) output by the delay device 41 is sent to the adaptive filter 42 and the filter tap coefficient arithmetic and control unit 44.
An example of the configuration of the adaptive filter 42 is shown in FIG. 14.
The illustrated adaptive filter 42 comprises a shift register composed of (n−1) number of series-connected memory elements S1–Sn−1, n number of multipliers J1–Jn, and (n−1) number of adders K1–Kn−1, where n is the number of filter taps.
The successive signals r(t−τ) output by the delay device 41 are input to the shift register to be stored in the (n−1) number of memory elements S1–Sn−1 in time sequence. The signals stored in the (n−1) number of memory elements S1–Sn−1 are sequentially shifted to the following memory elements.
The sequence u(t) of the input signals r(t−τ) in the shift register is represented by Equation (1). The symbol u(t) represents a vector.
In this specification, all symbols used to represent signals and the like are scalar unless explicitly stated to be vectors or sequences.u(t)={r1, r2, r3, . . . , rn}rx=r(t−τ−x+1)  (Eq. 1)
Signal r1 is a signal input to the shift register at a given time and output to the multiplier J1 without passing through any of the memory elements S1–Sn−1. Signals r2–rn are signals output from the memory elements S1–Sn−1 to the associated multipliers J2–Jn at their respective times.
In addition to being supplied with the associated signals r1–rn, the multipliers J1–Jn are also input with tap coefficient control signals h1–hn from the filter tap coefficient arithmetic and control unit 44 (explained later). The multipliers J1–Jn multiply the two input signals (i.e., weight the signals r1–rn with the tap coefficient control signals h1–hn) and output the multiplication results to adders K1–Kn−1.
The filter tap coefficient sequence h(t) output by the filter tap coefficient arithmetic and control unit 44 is represented by Equation 2, where h(t) is a vector.h(t)={h1, h2, h3, . . . , hn}  (Eq. 2)
The multiplication results output by the multipliers J1–Jn are summed by the adders K1–Kn−1, and the summing result is output to the adaptive filter 42. As explained further below, the filter tap coefficient arithmetic and control unit 44 sequentially updates the filter tap coefficient sequence h(t) so that the summing result is made the same signal as the interference signal contained in the received signal.
Specifically, the signal FM(t) output by the adaptive filter 42 (i.e., the summing result) is represented by Equation 3, where Σ means sum.FM(t)=h(t)*u(t)=Σ(hi*ri)(i=1, 2, . . . , n)  (Eq. 3)
In this specification, the symbol “*” means that the terms positioned before and after the symbol “*” are multiplied. In the particular case where two vectors are multiplied, it means a computation for calculating the inner product of the two vectors.
As explained earlier, the adaptive filter 42 is responsive to the tap coefficient control signal from the filter tap coefficient arithmetic and control unit 44 for extracting the aforesaid interference signal from the input signal r(t−τ) and outputting it to the subtractor 43 as the extracted interference wave signal FM(t).
The subtractor 43 receives the undelayed input signal r(t) and the output signal FM(t) from the adaptive filter 42, subtracts the output signal FM(t) from the input signal r(t), and outputs a substraction result e(t).
The substraction result e(t), the output signal of this interference signal removal system, is represented by Equation (4).e(t)=r(t)−FM(t)  (Eq. 4)
In this interference signal removal system, since the tap coefficient control signal from the filter tap coefficient arithmetic and control unit 44 (explained later) is sequentially updated, the extracted interference wave signal FM(t) is the same as the interference signal in the received signal. The substraction result e(t) is therefore the received signal removed of the interference signal, i.e., is the CDMA spread signal (ideally the spread signal only).
The filter tap coefficient arithmetic and control unit 44 receives the signal r(t−τ) output by the delay device 41 and the signal e(t) output by the subtractor 43, uses these signals to calculate a tap coefficient control signal that makes the signal FM(t) output by the adaptive filter 42 identical to the interference signal component, and outputs the calculated tap coefficient control signal to the adaptive filter 42.
The filter tap coefficient arithmetic and control unit 44 of the interference signal removal system uses an LMS (Least Mean Square), RLS (Recursive Least Square) or other such algorithm to calculate the tap coefficient control signal. Use of an LMS algorithm and an RLS algorthim will now be explained in the order mentioned.
The LMS general equation will be explained first.
The LMS update equation is expressed generally by Equation 5.h(t+1)=h(t)+μ*e(t)*u(t)  (Eq. 5)
Here, h(t) is the filter tap coefficient sequence at time t, and μ is a step size parameter (a coefficient related to convergence time and accuracy (weighting coefficient)), e(t) is the error signal at time t, and u(t) is the input signal sequence at time t.
The error signal e(t) is expressed generally by Equation 6.e(t)=d(t)−u(t)*h(t)  (Eq. 6)
For d(t), which is what is known as an ordinary unique word or training signal, there is used a known signal defined beforehand on the transmit side and the receive side. In the computation algorithm using Equations 5 and 6, the error signal e(t) is converged on 0 by sequentially updating the filter tap coefficient sequence.
Application of the LMS algorithm to the foregoing interference signal removal system will now be explained.
When Equation 5 is applied to the interference signal removal system, h(t) is the filter tap coefficient sequence output from the filter tap coefficient arithmetic and control unit 44 to the adaptive filter 42, and u(t) is signal sequence output from the delay device 41 to the filter tap coefficient arithmetic and control unit 44 (represented by Equation 1).
The interference signal removal system uses the signal output by the subtractor 43 (represented by Equation 4) as the error signal e(t). This is a characterizing feature of this interference signal removal circuit, and the processing differs from that by the ordinary LMS algorithm.
First, assume that the delay device 41 is not provided. In this case, since the computation algorithm moves the error signal e(t) toward 0, the signal e(t) output by the subtractor 43 would converge on 0 so that a filter tap coefficient sequence h(t) would be generated that removes the received signal of not only the interference signal but also the CDMA spread signal.
Since the interference signal removal system is in fact equipped with the delay device 41, however, there is a time difference equal to the delay time τ between the signal r(t−τ) input to the filter tap coefficient arithmetic and control unit 44 from the delay device 41 and the signal e(t) input to the filter tap coefficient arithmetic and control unit 44 through the subtractor 43.
The CDMA spread signal r(t) and the spread signal r(t−τ) delayed one or more chip times relative thereto are uncorrelated signals. Therefore, when it is attempted to converge the error signal e(t) on 0 in accordance with the foregoing computation algorithm, error e(t) remains because the spread signal component of u(t) is uncorrelated with r(t). In other words, in Equation 4, since successive summing of the input signal sequence u(t) makes the effect of the interference signal component locally 0, the spread signal component is not removed but remains as error e(t). On the other hand, since the interference signal component, which varies more gradually than the chip data, has correlation even if delayed on the order of several chip times, a filter tap coefficient sequence h(t) enabling removal of only the interference signal component from the received signal can be generated.
In other words, when the aforesaid computation algorithm is applied to the interference signal removal system, the filter tap coefficient sequence h(t) can be generated so as to leave the correlated component of u(t) and e(t) (i.e., the interference signal component) in the signal output by the adaptive filter 42 but not to leave the uncorrelated portion (i.e., the spread signal component) in the signal output by the adaptive filter 42.
Such a computation algorithm enables the adaptive filter 42 of the interference signal removal system to extract and output to the subtractor 43 only the interference signal component of the received signal, so that the subtractor 43 can output a signal obtained by removing only the interference signal component from the received signal (i.e., the CDMA spread signal).
The interference signal removal system shown in FIG. 13 thus utilizes the characteristics of the spread signal to process a received signal containing a broadband spread signal produced by CDMA spread-modulation and a narrow-band interference signal so as to adaptively remove the interference signal. It therefore prevents degradation and enhances the quality of reception.
While FIG. 13 shows a configuration in which the signal output by the subtractor 43 is not delayed, the same effect as the foregoing can also be obtained by adopting instead a configuration that, as shown in FIG. 15, delays the received signal input to a subtractor 53 using a delay device 51 and does not delay the received signal input to a filter tap coefficient arithmetic and control unit 54. The configuration of FIG. 15 is substantially the same as that of FIG. 13 aside from the provision of the delay device 51 on the side of the subtractor 53.
It is also possible to obtain a similar interference removal effect using an algorithm other than the foregoing LMS algorithm. A specific example of an update type when using an RLS algorithm in the configuration of FIG. 13 will now be explained. For convenience of explanation, parameters corresponding to the foregoing u(t), h(t), e(t), d(t) and r(t) will be assigned the same reference symbols.
For example, an n-row by 1-column vector consisting of the same components as u(t) represented by Equation 1 is defined as input sequence u(t), and an n-row by 1-column vector that, like h(t) represented by Equation 2, consists of n-number of filter tap coefficients is defined as filter tap coefficient sequence h(t).
Further, corresponding to the error signal e(t) represented by Equation 6, there is defined the RLS error signal e(t) represented by Equation 7 below, in which uT(t) is u(t) transposed.e(t)=d(t)−uT(t)*h(t)  (Eq. 7)
In the present example, the received signal r(t) input to the subtractor 43 is used as d(t), and uT(t)*h(t) in Equation 7 corresponds to the extracted interference wave signal output by the adaptive filter 42. In other words, as in the case of using the aforesaid LMS algorithm, the signal output by the adaptive filter 42 is used as the error signal e(t) represented by Equation 7. This is a characterizing feature of the present example. Similarly to when the LMS algorithm is used, error signal e(t) converges on 0 when the delay device 41 is not provided.
Using, for example, a coefficient error correlation matrix P(t) that is an n-row by n-column matrix and a gain vector k(t) that is an n-row by 1-column matrix, the RLS update equations can be written as the following Equations 8 to 10.h(t)=h(t−1)+k(t)*e(t)  (Eq. 8)k(t)={P(t−1)*u(t)}/{1+uT(t)*P(t−1)*u(t)}  (Eq. 9)P(t)=P(t−1)−k(t)*uT(t)*P(t−1)  (Eq. 10)
As the initial value h(0) of the filter tap coefficient sequence h(t) is used, for example, the zero vector represented by Equation 11, and as the initial value P(0) of the coefficient error correlation matrix P(t) is used, for example, a matrix wherein, as shown in Equation 12, all diagonal elements whose row number and column number coincide are a positive real number c and all other elements are 0 is used. hT(0) is h(0) transposed. Further, I in Equation 12 is an n-row by n-column matrix wherein all diagonal elements whose row number and column number coincide are 1 and all other elements are 0.hT(0)={0, 0, 0, . . . , 0}  (Eq. 11)
                              P          ⁡                      (            0            )                          =                              c            *            I                    =                      (                                                            c                                                  0                                                  .                                                  .                                                  0                                                                              0                                                  .                                                  .                                                  .                                                  .                                                                              .                                                  .                                                  c                                                  .                                                  .                                                                              .                                                  .                                                  .                                                  .                                                  0                                                                              0                                                  .                                                  .                                                  0                                                  c                                                      )                                              (                  Eq          .                                          ⁢          12                )            
As a result of the filter tap coefficient arithmetic and control unit 44 sequentially updating the filter tap coefficient sequence h(t) in accordance with the RLS update equations shown above, the signal output by the adaptive filter 42 can be brought progressively closer to the actual interference signal component, similarly to when the LMS algorithm is used, whereby a received signal containing a CDMA spread-modulated broadband signal and a narrow-band interference signal can be removed of the interference signal.
FIG. 16 shows an interference signal removal system that is supplied with the I component and Q component of a received signal containing a CDMA signal (desired signal) and an FM signal (interference signal) and removes the FM signal from the I component rI(t) and the Q component rQ(t). In this interference signal removal system, removal of the interference signal from the I component and the Q component of the received signal containing the CDMA spread-modulated signal and the interference signal is effected by imparting a time difference of one or more spreading code chip times between two signals into which the I component is split and two signals into which the Q component is split by time difference imparting means 61a, 61b, sending a received signal composed of one I component and Q component imparted with the time difference and a received signal composed of the other I component and Q component imparted with the time difference to extraction means 62a, 62b, 63a, 63b that extract from a signal component having correlation between the two signals an I component and a Q component of the interference signal as an interference signal component, and sending the I component and the Q component of the extracted interference signal component to a removal means 64a, 64b, 65a and 65b that use them to remove the I component of the interference signal component from the received signal and the Q component of the interference signal component from the received signal.
More specifically, the illustrated interference signal removal system comprises a delay device 61a for delaying an I phase signal (I component) orthogonally detected from the received signal, a delay device 61b for delaying a Q phase signal (Q component) orthogonally detected from the received signal, four adaptive filters 62a, 62b, 63a, 63b for extracting interference signal components from the delayed I component and Q component in accordance with a tap coefficient control signal from a filter tap coefficient arithmetic and control unit 66 (explained later), an adder 64a for summing the I component of the interference signal component, adder 64b for summing the Q component of the interference signal component, a subtractor 65a for removing the I component of the interference signal component from the I component of the received signal, a subtractor 65b for removing the Q component of the interference signal component from the Q component of the received signal, and the filter tap coefficient arithmetic and control unit 66 for outputting to the adaptive filters 62a, 62b, 63a, 63b the tap coefficient control signal based on the output signals from the subtractors 65a, 65b and the I component and Q component of the delayed received signal.
The configuration and operation of the illustrated circuit will now be explained.
The circuit is supplied with an I component rI(t) and a Q component rQ(t) orthogonally detected from a signal received by a receiver. The received signals rI(t), rQ(t) include a broadband spread signal, e.g. a CDMA spread-modulated signal, and a narrow-band communication signal constituting an interference signal (e.g., an FM-modulated signal). Similarly to what was explained earlier with reference to FIG. 13, the symbol t represents time and, in the present case, is an integer discrete value representing the minimum unit of one sample time.
The I component rI(t) is first split into two signals, one of which is input to the delay device 61a and the other of which is input to the subtractor 65a. Similarly, the Q component rQ(t) is first split into two signals, one of which is input to the delay device 61b and the other of which is input to the subtractor 65b. 
Similarly to the delay device 41 shown in FIG. 13, the delay devices 61a, 61b function to delay the input signals by one or more spreading code chip time widths. Both delay devices 61a, 61b impart the same delay time. Similarly to what was explained with reference to FIG. 13, the I component signal output by the delay device 61a is represented by rI(t−τ) and the Q component signal output by the delay device 61b is represented by rQ(t−τ), where τ is the delay time imparted by the delay devices 61a, 61b. 
The signal rI(t−τ) output by the delay device 61a is sent to the two adaptive filters 62a, 63a and the filter tap coefficient arithmetic and control unit 66. The signal rQ(t−τ) output by the delay device 61b is sent to the two adaptive filters 62b, 63b and the filter tap coefficient arithmetic and control unit 66.
The adaptive filters 62a, 62b, 63a, 63b can be of the same configuration as explained earlier with reference to FIG. 14. Four adaptive filters 62a, 62b, 63a, 63b are provided for carrying out the I-phase and Q-phase complex computations because the I component and the Q component of the interference signal component are contained in both the I component and the Q component of the received signal. Further, in this example, two types of filter tap coefficient sequences are used: an I-phase filter tap coefficient sequence hI(t) and a Q-phase filter tap coefficient sequence hQ(t). hI(t) and hQ(t) are vectors.
In this example, the filter tap coefficient arithmetic and control unit 66 (explained later) generates filter tap coefficient sequences hI(t), hQ(t) that enable the adaptive filter 62a to extract the I component of the interference signal component from the I component rI(t−τ) of the input received signal, the adaptive filter 63a to extract the Q component of the interference signal component from the I component rI(t−τ) of the input received signal, the adaptive filter 62b to extract the Q component of the interference signal component from the Q component rQ(t−τ) of the input received signal, and the adaptive filter 63b to extract the I component of the interference signal component from the Q component rQ(t−τ) of the input received signal.
The adder 64a sums the signals output by the adaptive filters 62a, 63b and outputs the result to the subtractor 65a. The summing result output to the subtractor 65a is the interference signal component of the I component of the received signal (i.e., the I component of the interference signal) FMI(t). In the present example, the adder 64a reverses the sign of the signal output by one adaptive filter (adaptive filter 63b) when carrying out the addition. If sign reversal is conducted by, for example, the adaptive filter 63b or the filter tap coefficient arithmetic and control unit 66 (explained later), however, the aforesaid sign reversal at the adaptive filter 62a is unnecessary.
The adder 64b sums the signals output by the adaptive filters 62b, 63a and outputs the result to the subtractor 65b. The summing result output to the subtractor 65b is the interference signal component of the Q component of the received signal (i.e., the Q component of the interference signal) FMQ(t).
The I component FMI(t) of the interference signal component output by the adder 64a is represented by Equation 13 and the Q component FMQ(t) of the interference signal component output by the adder 64b is represented by Equation 14. uI(t) and uQ(t) in Equations 13 and 14 are vectors and correspond to the I component and Q component of u(t) represented by Equation 1 in the explanation made earlier with reference to FIG. 13.FMI(t)={hI(t)*uI(t)}+{−hQ(t)*uQ(t)}  (Eq. 13)FMQ(t)={hI(t)*uQ(t)}+{hQ(t)*uI(t)}  (Eq. 14)
The subtractor 65a receives the undelayed I-component input signal rI(t) and the output signal FMI(t) from the adder 64a, subtracts the output signal FMI(t) from the input signal rI(t), and outputs a substraction result eI(t).
Similarly, the subtractor 65b receives the undelayed Q-component input signal rQ(t) and the output signal FMQ(t) from the adder 64b, subtracts the output signal FMQ(t) from the input signal rQ(t), and outputs a substraction result eQ(t).
The substraction results eI(t), eQ(t) are the output signals of this interference signal removal system.
In this interference signal removal system, since the tap coefficient control signal from the filter tap coefficient arithmetic and control unit 66 (explained later) is sequentially updated, the extracted I-component and Q-component interference wave signals FMI(t), FMQ(t) are the same as the I-component and Q-component interference signals in the received signal. The substraction results eI(t), eQ(t) are therefore the received signal from whose I component and Q component the interference signal has been removed, i.e., is the CDMA spread signal (ideally the spread signal only).
The filter tap coefficient arithmetic and control unit 66 receives the signals rI(t−τ), rQ(t−τ) output by the two delay devices 61a, 61b and the signals eI(t), eQ(t) output by the two subtractors 65a, 65b, uses these signals to calculate tap coefficient control signals that make the signals output by the adaptive filters 62a, 62b, 63a, 63b identical to the interference signal components, and outputs the calculated tap coefficient control signal to the adaptive filters 62a, 62b, 63a, 63b. In the present example, the two adaptive filters 62a, 62b output identical tap coefficient control signals, and the other two adaptive filters 63a, 63b output identical tap coefficient control signals, thereby setting them so as to generate the interference signal component FMI(t), FMQ(t) represented by Equations 13 and 14.
The filter tap coefficient arithmetic and control unit 66 of the interference signal removal system calculates the tap coefficient control signals using an algorithm for complex LMS algorithm such as explained with reference to FIG. 13. The LMS update equations of this algorithm are expressed by Equations 15 and 16.hI(t+1)=hI(t)+μ*(eI(t)*uI(t)+eQ(t)*uQ(t))  (Eq. 15)hQ(t+1)=hQ(t)+μ*(eQ(t)*uI(t)+eI(t)*uQ(t))  (Eq. 16)
Here, hI(t) and hQ(t) are the filter tap coefficient sequences at time t, and μ is a step size parameter (a coefficient related to convergence time and accuracy) and uI(t), uQ(t) are input signal sequences in the shift registers of the adaptive filters 62a, 63a and the shift registers of the adaptive filters 62b, 63b, respectively. Similarly to what was explained with reference to FIG. 13, moreover, as eI(t), eQ(t) there are used the signals output by the subtractor 65a and the subtractor 65b, respectively. As stated earlier, uI(t) and uQ(t) are vectors.
Similarly to what was explained with respect to FIG. 13, since in this interference signal removal system the computation algorithm sequentially updates the filter tap coefficient sequences hI(t), hQ(t), filter tap coefficient sequences hI(t), hQ(t) can be generated that are capable of removing the interference signal components, which have relatively strong correlation, without removing the spread signal component, which has uncorrelation.
Since the filter tap coefficient sequences hI(t), hQ(t) are calculated taking both the I component and the Q component into account, moreover, the accuracy of the interference removal is improved.
The interference signal removal system shown in FIG. 16 thus utilizes the characteristics of the spread signal to process a received signal containing a broadband spread signal produced by CDMA spread-modulation and a narrow-band interference signal so as to remove the interference signal from the I component and Q component thereof. It therefore prevents degradation and enhances the quality of reception.
While FIG. 16, like FIG. 13 referred to earlier, shows a configuration in which the signals output by the subtractors 65a, 65b are not delayed, the same effect as the foregoing can also be obtained by instead adopting a configuration that, as shown in FIG. 17, delays the received signal input to subtractors 75a, 75b using delay devices 71a, 71b and does not delay the received signal input to adaptive filters 72a, 72b, 73a, 73b and a filter tap coefficient arithmetic and control unit 76. The configuration of FIG. 17 is substantially the same as that of FIG. 16 aside from the provision of the delay devices 71a, 71b on the side of the subtractors 74a, 74b in addition to the constituents set out in the foregoing.
Also similarly to what was explained with reference to FIG. 13, it is possible to obtain a similar interference removal effect using an algorithm other than the foregoing LMS algorithm. An example using an RLS algorithm for complex computation in the configuration of FIG. 16 will now be explained. For convenience of explanation, parameters corresponding to the foregoing uI(t), uQ(t), hI(t), hQ(t), eI(t), eQ(t), rI(t) and rQ(t) will be assigned the same reference symbols.
In the RLS algorithm for complex computation, u(t), h(t), e(t), k(t), P(t) and all other parameters indicated in Equations 7 to 10 consist of complex number terms. Defining γ and ω as real numbers and using j as the symbol for the imaginary part, an arbitrary complex number term is represented as (γ+jω).
With the RLS algorithm for complex computation, sequential update processing like that explained with reference to FIG. 13 can be realized in complex computation by separating the real number part and the imaginary number part and using them as the I-component parameter and the Q-component parameter, respectively.
Specifically, in the case of the present example, the real number part of u(t) is defined as uI(t), the imaginary number part thereof as uQ(t), the real number part of h(t) is defined as hI(t) and the imaginary number part thereof as hQ(t), the real number part of eI(t) is defined as eI(t) and the imaginary number part thereof as eQ(t), and so forth, and processing is conducted for removing the interference signal components from the I component rI(t) and Q component rQ(t) of the received signal.
Thus, similarly to when an LMS algorithm for complex computation is used, use of an RLS algorithm for complex computation in the foregoing manner also enables removal of the interference signal from the I component and the Q component of a received signal containing a CDMA spread-modulated broadband signal and a narrow-band interference signal.
An explanation will now be given regarding how the conventional interference signal removal system such as that shown in FIG. 10 processes a received signal to remove a periodic TDMA signal received as interference signal.
FIG. 18(a) shows an example of the time course of the received power of a received signal when a periodic TDMA signal is superimposed as an interference signal on a broadband signal such as a CDMA signal received as the desired signal. In the drawing, time (t) is represented on the horizontal axis and received power on the vertical axis. FIG. 18(a) shows an example in which interference signal power that varies periodically, i.e., repeatedly appears every regular interval T, is superimposed on broadband signal power that is substantial constant over time. From FIG. 18(a) it can be seen that when a periodic TDMA signal is received as interference, the received power of the received signal varies dependently thereon at the same period T. FIG. 18(a) shows a case in which interference signal power of two magnitudes is superimposed on the broadband signal power.
Moreover, interference from TDMA signals is not necessarily limited to a single frequency and may be received at multiple frequencies.
FIG. 18(b) shows an example of the time course of received signal frequency components when a broadband desired signal receives interference from TDMA signals in two different frequency bands. The horizontal axis of the chart represents time (t) and the vertical axis frequency. On the horizontal axis representing time elapse, the time T of each period is divided into 8 unit time periods T1–T8 of equal time width ΔT.
More specifically, FIG. 18(b) shows an example in which a TDMA signal of frequency f3 that repeatedly appears every time period T1 and a TDMA signal of frequency f2 that repeatedly appears every time period T4 and time period T8 interfere with a broadband signal spanning frequency f1 to frequency f4 (f1>f2>f3>f4). The places where interference occurs are colored black. The interference signals usually have a bandwidth. The expressions “interference signal of frequency f3” and “interference signal of frequency f2” are therefore used to specify the interference signals by their center frequencies f3 and f2.
In order for a conventional adaptive filter type interference signal removal system such as shown in FIG. 10 to deal with interference signals that exhibit the intense power change of a TDMA signal, the update rate of the control coefficient h(t+1) used for interference signal estimation and extraction must be accelerated during the interference signal estimation conducted by the sequential update processing of the interference signal estimator 31.
FIG. 18(c) shows an example of the time course of the power of the interference-reduced received signal obtained by removing the TDMA signal (the interference signal) from the received signal shown in FIG. 18(a) using a conventional interference signal removal system operated at an accelerated control coefficient h(t+1). The horizontal axis of the chart represents time (t) and the vertical axis represents received signal power after interference signal removal. For convenience of explanation, FIG. 18(c) shows power of the received signal after removal of the interference signals by a solid curve and the interference signal before interference signal removal by a dotted curve.
FIG. 18(d) shows an example of the time course of the frequency components of the interference-reduced received signal obtained by removing the TDMA signals (the interference signals) from the received signal shown in FIG. 18(b) using a conventional interference signal removal system operated at an accelerated control coefficient h(t+1). The horizontal axis of the chart represents time (t) and the vertical axis frequency. Places where interference signal components remain are represented in shades of gray. The shading approaches white with decreasing amount of remaining interference signal after interference removal.
As shown in FIGS. 18(c) and 18(d), in the case of interference signals that vary intensely like TDMA signals, the conventional interference signal removal system cannot, merely by speeding up the update rate of the control coefficient h(t+1), thoroughly remove interference signals, because sufficient time for interference signal estimation cannot be obtained at the beginning of interference signal reception. Moreover, a configuration that updates the control coefficient h(t+1) for interference signal estimation and extraction at high speed is not practical from the aspect of system feasibility. Although a method that does not sequentially update the control coefficient h(t+1) and operates the filter using a constant control coefficient h(t+1) even when no interference signal is present is conceivable, this method is disadvantageous because the effect of the interference signals on the received signal increases and reception quality degenerates when interference is received in multiple frequency bands.
Some conventional techniques for interference signal removal will now be set out for reference purposes.
In the “Method and device for conducting simultaneous broadband and narrow-band communication” taught by Patent Kohyo (Japanese National Publication of Translated Version) No. 9-507734 a high-intensity signal component of a communication signal is detected by FFT (Fast Fourier Transformation), the detected signal component is removed from the received signal by a frequency notch filter, and the received signal after the removal is subjected to iFFT (inverse FFT).
In the “Interference wave cancellation device” taught by Japanese Unexamined Patent Publication No. 2000-174645, an interference wave cancellation signal is generated by wave-filtering a received signal with a narrow-band BPF (Band Pass Filter) exhibiting a substantially flat phase characteristic in a specific frequency band of significant width.
In the “Narrow-band interference wave restricting device and communication using the same” taught by Japanese Unexamined Patent Publication No. 2000-196497, an interference wave is restricted by using a frequency-selective level restricting circuit (e.g., a magnetostatic wave filter) to restrict the output level of a variable amplification factor amplifier, the power of a spread signal output by the frequency-selective level restricting circuit is detected by a power detector, and variable amplification factor is controlled to make the detected value constant.
In the “Noise removal device and noise removal method in code division multiple access communications” taught by Japanese Unexamined Patent Publication No. 2000-307468, a jamming wave carrier frequency including a jamming wave signal is detected from a received signal using a pair of receiver units and a jamming wave discriminating unit and the detected jamming wave carrier wave signal is removed from the received signal using a filter unit.
As is clear from the foregoing discussion of the prior art, conventional interference signal removal systems cannot achieve sufficient interference signal removal accuracy when removing interference signals that vary intensely like TDMA signals. Moreover, conventional interference signal removal systems have not given adequate attention to techniques for efficient removal of interference signals that appear repeatedly in the manner of TDMA signals.
This invention was accomplished in light of these circumstances and has as an object to provide an interference signal removal system that enables efficient interference signal removal in cases where, for example, an interference signal superimposed on a desired signal and periodically varying in received power level is removed from the received signal. Another object of the invention is to provide an interference signal removal system that enables accurate interference signal removal even in a case of removing from a received signal a TDMA signal or other such intensely varying signal.